# Inductance

(Redirected from Self inductance)

In electromagnetism and electronics, inductance is the property of an electrical conductor by which a change in current through it induces an electromotive force in both the conductor itself[1] and in any nearby conductors by mutual inductance.[1]

These effects are derived from two fundamental observations of physics: a steady current creates a steady magnetic field described by Oersted's law,[2] and a time-varying magnetic field induces an electromotive force (EMF) in nearby conductors, which is described by Faraday's law of induction.[3] According to Lenz's law,[4] a changing electric current through a circuit that contains inductance induces a proportional voltage, which opposes the change in current (self-inductance). The varying field in this circuit may also induce an EMF in neighbouring circuits (mutual inductance).

The term inductance was coined by Oliver Heaviside in 1886.[5] It is customary to use the symbol L for inductance, in honour of the physicist Heinrich Lenz.[6][7] In the SI system, the measurement unit for inductance is the henry, with the unit symbol H, named in honor of Joseph Henry, who discovered inductance independently of, but not before, Faraday.[8]

## Circuit analysis

An electronic component that is intended to add inductance to a circuit is called an inductor. Inductors are typically manufactured from coils of wire. This design delivers two desired properties, a concentration of the magnetic field into a small physical space and a linking of the magnetic field into the circuit multiple times.[citation needed]

The relationship between the self-inductance, L, of an electrical circuit, the voltage, v(t), and the current, i(t), through the circuit is

${\displaystyle \displaystyle v(t)=L\,{\frac {di(t)}{dt}}}$.

A voltage is induced across an inductor (back EMF), that is equal to the product of the inductor's inductance and the rate of change of current through the inductor.

All circuits have, in practice, some inductance, which may have beneficial or detrimental effects. For a tuned circuit, inductance is used to provide a frequency-selective circuit. Practical inductors may be used to provide filtering, or energy storage, in a given network. The inductance per unit length of a transmission line is one of the properties that determines its characteristic impedance; balancing the inductance and capacitance of cables is important for distortion-free telegraphy and telephony. The inductance of long AC power transmission lines affects the power capacity of the line. Sensitive circuits, such as microphone and computer network cables, may utilize special cabling construction, limiting the inductive coupling between circuits.

The generalization to the case of K electrical circuits with currents, {im}, and voltages, {vm}, reads

${\displaystyle \displaystyle v_{m}=\sum \limits _{n=1}^{K}L_{m,n}\,{\frac {di_{n}}{dt}}.}$

Here, inductance L is a symmetric matrix. The diagonal coefficients Lm,m are called coefficients of self-inductance, the off-diagonal elements are called coefficients of mutual inductance. The coefficients of inductance are constant, as long as no magnetizable material with nonlinear characteristics is involved. This is a direct consequence of the linearity of Maxwell's equations in the fields and the current density. The coefficients of inductance become functions of the currents in the nonlinear case.

## Derivation from Faraday's law of inductance

The inductance equations above are a consequence of Maxwell's equations. There is a straightforward derivation in the important case of electrical circuits consisting of thin wires.

In a system of K wire loops, each with one or several wire turns, the flux linkage of loop m, λm, is given by

${\displaystyle \displaystyle \lambda _{m}=N_{m}\Phi _{m}=\sum \limits _{n=1}^{K}L_{m,n}i_{n}.}$

Here Nm denotes the number of turns in loop m; Φm, the magnetic flux through loop m; and Lm,n are some constants. This equation follows from Ampere's law - magnetic fields and fluxes are linear functions of the currents. By Faraday's law of induction, we have

${\displaystyle \displaystyle v_{m}={\frac {d\lambda _{m}}{dt}}=N_{m}{\frac {d\Phi _{m}}{dt}}=\sum \limits _{n=1}^{K}L_{m,n}{\frac {di_{n}}{dt}},}$

where vm denotes the voltage induced in circuit m. This agrees with the definition of inductance above if the coefficients Lm,n are identified with the coefficients of inductance. Because the total currents Nnin contribute to Φm it also follows that Lm,n is proportional to the product of turns NmNn.

## Inductance and magnetic field energy

Multiplying the equation for vm above with imdt and summing over m gives the energy transferred to the system in the time interval dt,

${\displaystyle \displaystyle \sum \limits _{m}^{K}i_{m}v_{m}dt=\sum \limits _{m,n=1}^{K}i_{m}L_{m,n}di_{n}{\overset {!}{=}}\sum \limits _{n=1}^{K}{\frac {\partial W\left(i\right)}{\partial i_{n}}}di_{n}.}$

This must agree with the change of the magnetic field energy, W, caused by the currents.[9] The integrability condition

${\displaystyle \displaystyle {\frac {\partial ^{2}W}{\partial i_{m}\partial i_{n}}}={\frac {\partial ^{2}W}{\partial i_{n}\partial i_{m}}}}$

requires Lm,n = Ln,m. The inductance matrix, Lm,n, thus is symmetric. The integral of the energy transfer is the magnetic field energy as a function of the currents,

${\displaystyle \displaystyle W\left(i\right)={\frac {1}{2}}\sum \limits _{m,n=1}^{K}i_{m}L_{m,n}i_{n}.}$

This equation also is a direct consequence of the linearity of Maxwell's equations. It is helpful to associate changing electric currents with a build-up or decrease of magnetic field energy. The corresponding energy transfer requires or generates a voltage. A mechanical analogy in the K = 1 case with magnetic field energy (1/2)Li2 is a body with mass M, velocity u and kinetic energy (1/2)Mu2. The rate of change of velocity (current) multiplied with mass (inductance) requires or generates a force (an electrical voltage).

## Coupled inductors and mutual inductance

The circuit diagram representation of mutually coupled inductors. The two vertical lines between the inductors indicate a solid core that the wires of the inductor are wrapped around. "n:m" shows the ratio between the number of windings of the left inductor to windings of the right inductor. This picture also shows the dot convention.

Mutual inductance occurs when the change in current in one inductor induces a voltage in another nearby inductor. It is important as the mechanism by which transformers work, but it can also cause unwanted coupling between conductors in a circuit.

The mutual inductance, M, is also a measure of the coupling between two inductors. The mutual inductance by circuit i on circuit j is given by the double integral Neumann formula, see calculation techniques

The mutual inductance also has the relationship:

${\displaystyle M_{21}=N_{1}N_{2}P_{21}\!}$

where

${\displaystyle M_{21}}$ is the mutual inductance, and the subscript specifies the relationship of the voltage induced in coil 2 due to the current in coil 1.
N1 is the number of turns in coil 1,
N2 is the number of turns in coil 2,
P21 is the permeance of the space occupied by the flux.

Once the mutual inductance, M, is determined, it can be used to predict the behavior of a circuit:

${\displaystyle v_{1}=L_{1}{\frac {di_{1}}{dt}}-M{\frac {di_{2}}{dt}}}$

where

v1 is the voltage across the inductor of interest,
L1 is the inductance of the inductor of interest,
di1/dt is the derivative, with respect to time, of the current through the inductor of interest,
di2/dt is the derivative, with respect to time, of the current through the inductor that is coupled to the first inductor, and
M is the mutual inductance.

The minus sign arises because of the sense the current i2 has been defined in the diagram. With both currents defined going into the dots the sign of M will be positive (the equation would read with a plus sign instead).[10]

### Matrix representation

The circuit can be described by any of the two-port network parameter matrix representations. The most direct are the z parameters, which are given by

${\displaystyle [\mathbf {z} ]=s{\begin{bmatrix}L_{1}\ M\\M\ L_{2}\end{bmatrix}}}$

where s is the complex frequency variable.

### Coupling coefficient

The coupling coefficient is the ratio of the open-circuit actual voltage ratio to the ratio that would obtain if all the flux coupled from one circuit to the other. The coupling coefficient is related to mutual inductance and self inductances in the following way. From the two simultaneous equations expressed in the 2-port matrix the open-circuit voltage ratio is found to be:

${\displaystyle {V_{2} \over V_{1}}({\text{open circuit}})={M \over L_{1}}}$

while the ratio if all the flux is coupled is the ratio of the turns, hence the ratio of the square root of the inductances

${\displaystyle {V_{2} \over V_{1}}({\text{max coupled}})={\sqrt {L_{2} \over L_{1}}}}$

thus,

${\displaystyle M=k{\sqrt {L_{1}L_{2}}}}$

where

k is the coupling coefficient,
L1 is the inductance of the first coil, and
L2 is the inductance of the second coil.

The coupling coefficient is a convenient way to specify the relationship between a certain orientation of inductors with arbitrary inductance. Most authors define the range as 0 ≤ k < 1, but some[11] define it as −1 < k < 1. Allowing negative values of k captures phase inversions of the coil connections and the direction of the windings.[12]

### Equivalent circuits

#### T-circuit

T equivalent circuit of mutually coupled inductors

Mutually coupled inductors can equivalently be represented by a T-circuit of inductors as shown. If the coupling is strong and the inductors are of unequal values then the series inductor on the step-down side may take on a negative value.

This can be analyzed as a two port network. With the output terminated with some arbitrary impedance, Z, the voltage gain, Av, is given by,

${\displaystyle A_{\mathrm {v} }={\frac {sMZ}{\,s^{2}L_{1}L_{2}-s^{2}M^{2}+sL_{1}Z\,}}={\frac {k}{\,s\left(1-k^{2}\right){\frac {\sqrt {L_{1}L_{2}}}{Z}}+{\sqrt {\frac {L_{1}}{L_{2}}}}\,}}}$

where k is the coupling constant and s is the complex frequency variable, as above. For tightly coupled inductors where k = 1 this reduces to

${\displaystyle A_{\mathrm {v} }={\sqrt {L_{2} \over L_{1}}}}$

which is independent of the load impedance. If the inductors are wound on the same core and with the same geometry, then this expression is equal to the turns ratio of the two inductors because inductance is proportional to the square of turns ratio.

The input impedance of the network is given by,

${\displaystyle Z_{\mathrm {in} }={\frac {s^{2}L_{1}L_{2}-s^{2}M^{2}+sL_{1}Z}{sL_{2}+Z}}={\frac {L_{1}}{L_{2}}}\,Z\,{\biggl (}{\frac {1}{1+\left({\frac {Z}{\,sL_{2}\,}}\right)}}{\biggr )}{\Biggl (}1+{\frac {\left(1-k^{2}\right)}{\left({\frac {Z}{\,sL_{2}\,}}\right)}}{\Biggr )}}$

For k = 1 this reduces to

${\displaystyle Z_{\mathrm {in} }={\frac {sL_{1}Z}{sL_{2}+Z}}={\frac {L_{1}}{L_{2}}}\,Z\,{\biggl (}{\frac {1}{1+\left({\frac {Z}{\,sL_{2}\,}}\right)}}{\biggr )}}$

Thus, the current gain, Ai is not independent of load unless the further condition

${\displaystyle |sL_{2}|\gg |Z|}$

is met, in which case,

${\displaystyle Z_{\mathrm {in} }\approx {L_{1} \over L_{2}}Z}$

and

${\displaystyle A_{\mathrm {i} }\approx {\sqrt {L_{1} \over L_{2}}}={1 \over A_{\mathrm {v} }}}$

#### π-circuit

π equivalent circuit of coupled inductors

Alternatively, coupled inductors can be modelled using a π equivalent circuit as shown for two inductors. While the circuit is more complicated than a T-circuit, it can be generalized [13] to higher-order circuits, that is circuits consisting of more than two coupled inductors. Also, as circuit elements Ls, Lp have physical meaning, denoting respectively magnetic reluctance of a coupling path and magnetic reluctance of a leakage path, electric currents through these devices correspond to normalized magnetic fluxes in these paths. Ideal transformers at each port model self-inductances of each coupled inductor and normalize them to 1H.

Circuit element values can be calculated from coupling coefficients with

${\displaystyle L_{S_{ij}}={\dfrac {\det(\mathbf {K} )}{-\mathbf {C} _{ij}}}}$
${\displaystyle L_{P_{i}}={\dfrac {\det(\mathbf {K} )}{\sum _{j=1}^{N}\mathbf {C} _{ij}}}}$

where

${\displaystyle \mathbf {K} ={\begin{bmatrix}1&k_{12}&\cdots &k_{1N}\\k_{12}&1&\cdots &k_{2N}\\\vdots &\vdots &\ddots &\vdots \\k_{1N}&k_{2N}&\cdots &1\end{bmatrix}};\mathbf {C} _{ij}=(-1)^{i+j}\,\mathbf {M} _{ij}}$

For two coupled inductors, these formulas simplify to:

${\displaystyle L_{S_{12}}={\dfrac {-k_{12}^{2}+1}{k_{12}}};L_{P_{1}}=L_{P_{2}}\!=\!k_{12}+1}$

and for three coupled inductors (for brevity shown only for Ls12 and Lp1):

${\displaystyle L_{S_{12}}={\frac {2\,k_{12}\,k_{13}\,k_{23}-k_{12}^{2}-k_{13}^{2}-k_{23}^{2}+1}{k_{13}\,k_{23}-k_{12}}};L_{P_{1}}={\frac {2\,k_{12}\,k_{13}\,k_{23}-k_{12}^{2}-k_{13}^{2}-k_{23}^{2}+1}{k_{12}\,k_{23}+k_{13}\,k_{23}-k_{23}^{2}-k_{12}-k_{13}+1}}}$

### Tuned transformer

When either side of the transformer is a tuned circuit, the amount of mutual inductance between the two windings, together with the Q factor of the circuit, determine the shape of the frequency response curve. The tuned circuit together with the transformer load form an RLC circuit with a definite peak in the frequency response. When both sides of the transformer are tuned, it is described as double-tuned. The coupling of double-tuned circuits is described as loose-, critical-, or over-coupled depending on the value of k. When two tuned circuits are loosely coupled through mutual inductance, the bandwidth will be narrow. As the amount of mutual inductance increases, the bandwidth continues to grow. When the mutual inductance is increased beyond a critical point, the peak in the response curve begins to drop, and the center frequency will be attenuated more strongly than its direct sidebands. This is known as overcoupling.

### Ideal transformers

When k = 1, the inductor is referred to as being closely coupled. If in addition, the self-inductances go to infinity, the inductor becomes an ideal transformer. In this case the voltages, currents, and number of turns can be related in the following way:

${\displaystyle V_{\text{s}}={\frac {N_{\text{s}}}{N_{\text{p}}}}V_{\text{p}}}$

where

Vs is the voltage across the secondary inductor,
Vp is the voltage across the primary inductor (the one connected to a power source),
Ns is the number of turns in the secondary inductor, and
Np is the number of turns in the primary inductor.

Conversely the current:

${\displaystyle I_{\text{s}}={\frac {N_{\text{p}}}{N_{\text{s}}}}I_{\text{p}}}$

where

Is is the current through the secondary inductor,
Ip is the current through the primary inductor (the one connected to a power source),
Ns is the number of turns in the secondary inductor, and
Np is the number of turns in the primary inductor.

The power through one inductor is the same as the power through the other. These equations neglect any forcing by current sources or voltage sources.

## Calculation techniques

In the most general case, inductance can be calculated from Maxwell's equations. Many important cases can be solved using simplifications. Where high frequency currents are considered, with skin effect, the surface current densities and magnetic field may be obtained by solving the Laplace equation. Where the conductors are thin wires, self-inductance still depends on the wire radius and the distribution of the current in the wire. This current distribution is approximately constant (on the surface or in the volume of the wire) for a wire radius much smaller than other length scales.

### Mutual inductance of two wire loops

The mutual inductance by a filamentary circuit ${\displaystyle m}$ on a filamentary circuit ${\displaystyle n}$ is given by the double integral Neumann formula[14]

${\displaystyle L_{m,n}={\frac {\mu _{0}}{4\pi }}\oint _{C_{m}}\oint _{C_{n}}{\frac {d\mathbf {x} _{m}\cdot d\mathbf {x} _{n}}{|\mathbf {x} _{m}-\mathbf {x} _{n}|}}.}$

The symbol μ0 denotes the magnetic constant (4π × 10−7 H/m), Cm and Cn are the curves spanned by the wires. See a derivation of this equation.

### Self-inductance of a wire loop

Formally, the self-inductance of a wire loop would be given by the above equation with ${\displaystyle m}$ = ${\displaystyle n}$. The problem, however, is that ${\displaystyle 1/|\mathbf {x} -\mathbf {x} '|}$ now becomes infinite, leading to a logarithmically divergent integral. This necessitates taking the finite wire radius ${\displaystyle a}$ and the distribution of the current in the wire into account. There remain the contribution from the integral over all points with ${\displaystyle |\mathbf {x} -\mathbf {x} '|}$ > ${\displaystyle a/2}$ and a correction term,[15]

${\displaystyle L=\left({\frac {\mu _{0}}{4\pi }}\oint _{C}\oint _{C'}{\frac {d\mathbf {x} \cdot d\mathbf {x} '}{|\mathbf {x} -\mathbf {x} '|}}\right)_{|\mathbf {x} -\mathbf {x} '|>{\frac {a}{2}}}+{\frac {\mu _{0}}{4\pi }}lY+O\left(\mu _{0}a\right).}$

Here ${\displaystyle a}$ and ${\displaystyle l}$ denote the radius and length of the wire, and ${\displaystyle Y}$ is a constant that depends on the distribution of the current in the wire: ${\displaystyle Y=0}$ when the current flows in the surface of the wire (skin effect), ${\displaystyle Y=1/2}$ when the current is homogeneous across the wire. The error ${\displaystyle O(\mu _{0}a)}$ is small when the wire is long compared to its radius.

### Method of images

In some cases, different current distributions generate the same magnetic field in some section of space. This fact may be used to relate self inductances (method of images). As an example, consider the two systems:

• A wire at distance ${\displaystyle d/2}$ in front of a perfectly conducting wall (which is the return)
• Two parallel wires at distance ${\displaystyle d}$, with opposite current

The magnetic field of the two systems coincides (in a half space). The magnetic field energy and the inductance of the second system thus are twice as large as that of the first system.

### Relation between inductance and capacitance

Inductance per length L' and capacitance per length C' are related to each other in the special case of transmission lines consisting of two parallel perfect conductors of arbitrary but constant cross section,[16]

${\displaystyle \displaystyle L'C'={\varepsilon \mu }.}$

Here ε and µ denote the dielectric constant and magnetic permeability of the medium that the conductors are embedded in. There is no electric and no magnetic field inside the conductors (complete skin effect, high frequency). Current flows down on one line and returns on the other. Signals will propagate along the transmission line at the speed of electromagnetic radiation in the non-conductive medium enveloping the conductors.

## Self-inductance of simple electrical circuits in air

The self-inductance of many types of electrical circuits can be given in closed form. Examples are listed in the table.

Inductance of simple electrical circuits in air
Type Inductance Comment
Single layer
solenoid[17]
{\displaystyle {\begin{aligned}&{\frac {\mu _{0}r^{2}N^{2}}{3l}}\left(-8w+4{\frac {\sqrt {1+m}}{m}}\left[K{\sqrt {\frac {m}{1+m}}}-\left(1-m\right)E{\sqrt {\frac {m}{1+m}}}\right]\right)\\={}&{\frac {\mu _{0}r^{2}N^{2}\pi }{l}}\left[1-{\frac {8w}{3\pi }}+\sum _{n=1}^{\infty }{\frac {{\left(2n\right)!}^{2}}{n!^{4}\left(n+1\right)\left(2n-1\right)2^{2n}}}\left(-1\right)^{n+1}w^{2n}\right]\\\approx {}&{\frac {\mu _{0}r^{2}N^{2}\pi }{l}}\left(1-{\frac {8w}{3\pi }}+{\frac {w^{2}}{2}}-{\frac {w^{4}}{4}}+{\frac {5w^{6}}{16}}-{\frac {35w^{8}}{64}}+\ldots \right){\text{for }}w\ll 1\\\approx {}&\mu _{0}rN^{2}\left(\left[1+{\frac {1}{32w^{2}}}+O\left({\frac {1}{w^{4}}}\right)\right]\ln(8w)-{\frac {1}{2}}+{\frac {1}{128w^{2}}}+O\left[{\frac {1}{w^{4}}}\right]\right){\text{for }}w\gg 1\end{aligned}}}
• ${\displaystyle N:}$ Number of turns
• ${\displaystyle r:}$ Radius
• ${\displaystyle l:}$ Length
• ${\displaystyle w:r/l}$
• ${\displaystyle m:4w^{2}}$
• ${\displaystyle E,K:}$ Elliptic integrals
Coaxial cable,
high frequency
${\displaystyle {\frac {\mu _{0}l}{2\pi }}\ln \left({\frac {a_{1}}{a}}\right)}$ ${\displaystyle a_{1}:}$ Outer radius
${\displaystyle a:}$ Inner radius
${\displaystyle l:}$ Length
Circular loop[18] ${\displaystyle \mu _{0}r\left[\ln \left({\frac {8r}{a}}\right)-2+{\frac {Y}{2}}+O\left({\frac {a^{2}}{r^{2}}}\right)\right]}$ ${\displaystyle r:}$ Loop radius
${\displaystyle a:}$ Wire radius
Rectangle[19] {\displaystyle {\begin{aligned}&{\frac {\mu _{0}}{\pi }}\left[b\ln \left({\frac {2b}{a}}\right)+d\ln \left({\frac {2d}{a}}\right)-\left(b+d\right)\left(2-{\frac {Y}{2}}\right)+2{\sqrt {b^{2}+d^{2}}}\right]\\-&{\frac {\mu _{0}}{\pi }}\left[b\operatorname {arsinh} \left({\frac {b}{d}}\right)+d\operatorname {arsinh} \left({\frac {d}{b}}\right)+O\left(a\right)\right]\end{aligned}}} ${\displaystyle b,d:}$ Border length
${\displaystyle d\gg a,b\gg a}$
${\displaystyle a:}$ Wire radius
Pair of parallel
wires
${\displaystyle {\frac {\mu _{0}l}{\pi }}\left[\ln \left({\frac {d}{a}}\right)+{\frac {Y}{2}}\right]}$ ${\displaystyle a:}$ Wire radius
${\displaystyle d:}$ Distance, ${\displaystyle d\geq 2a}$
${\displaystyle l:}$ Length of pair
Pair of parallel
wires, high
frequency
${\displaystyle {\frac {\mu _{0}l}{\pi }}\operatorname {arcosh} \left({\frac {d}{2a}}\right)={\frac {\mu _{0}l}{\pi }}\ln \left({\frac {d}{2a}}+{\sqrt {{\frac {d^{2}}{4a^{2}}}-1}}\right)}$ ${\displaystyle a:}$ Wire radius
${\displaystyle d:}$ Distance, ${\displaystyle d\geq 2a}$
${\displaystyle l}$: Length of pair
Wire parallel to
perfectly
conducting wall
${\displaystyle {\frac {\mu _{0}l}{2\pi }}\left[\ln \left({\frac {2d}{a}}\right)+{\frac {Y}{2}}\right]}$ ${\displaystyle a:}$ Wire radius
${\displaystyle d:}$ Distance, ${\displaystyle d\geq a}$
${\displaystyle l}$: Length
Wire parallel to
conducting wall,
high frequency
${\displaystyle {\frac {\mu _{0}l}{2\pi }}\operatorname {arcosh} \left({\frac {d}{a}}\right)={\frac {\mu _{0}l}{2\pi }}\ln \left({\frac {d}{a}}+{\sqrt {{\frac {d^{2}}{a^{2}}}-1}}\right)}$ ${\displaystyle a:}$ Wire radius
${\displaystyle d:}$ Distance, ${\displaystyle d\geq a}$
${\displaystyle l}$: Length

The symbol μ0 denotes the magnetic constant (4π×10−7 H/m) in SI units.

The purpose of the constant ${\displaystyle Y:}$ For high frequencies, the electric current flows in the conductor surface (skin effect) and, depending on the geometry, it is sometimes necessary to distinguish low and high frequency inductances. For high frequencies, ${\displaystyle Y=0}$ when the current is uniformly distributed over the surface of the wire (skin effect), ${\displaystyle Y=0.5}$ when the current is uniformly distributed over the cross section of the wire (very low frequencies). In the high frequency case, if conductors are within one wire diameter of each other, an additional screening current flows in their surfaces, and expressions containing ${\displaystyle Y}$ become invalid.

## Inductance with physical symmetry

### Inductance of a solenoid

A solenoid is a long, thin coil; i.e., a coil whose length is much greater than its diameter. Under these conditions, and without any magnetic material used, the magnetic flux density ${\displaystyle B}$ within the coil is practically constant and is given by

${\displaystyle \displaystyle B={\frac {\mu _{0}Ni}{l}}}$

where ${\displaystyle \mu _{0}}$ is the magnetic constant, ${\displaystyle N}$ the number of turns, ${\displaystyle i}$ the current and ${\displaystyle l}$ the length of the coil. Ignoring end effects, the total magnetic flux through the coil is obtained by multiplying the flux density ${\displaystyle B}$ by the cross-section area ${\displaystyle A}$:

${\displaystyle \displaystyle \Phi ={\frac {\mu _{0}NiA}{l}},}$

When this is combined with the definition of inductance,

${\displaystyle \displaystyle L={\frac {N\Phi }{i}}}$

it follows that the inductance of a solenoid is given by:

${\displaystyle \displaystyle L={\frac {\mu _{0}N^{2}A}{l}}.}$

A table of inductance for short solenoids of various diameter to length ratios has been calculated by Dellinger, Whittmore, and Ould.[20]

This, and the inductance of more complicated shapes, can be derived from Maxwell's equations. For rigid air-core coils, inductance is a function of coil geometry and number of turns, and is independent of current.

Similar analysis applies to a solenoid with a magnetic core, but only if the length of the coil is much greater than the product of the relative permeability of the magnetic core and the diameter. That limits the simple analysis to low-permeability cores, or extremely long thin solenoids. Although rarely useful, the equations are,

${\displaystyle \displaystyle B={\frac {\mu _{0}\mu _{r}Ni}{l}}}$

where ${\displaystyle \mu _{r}}$ the relative permeability of the material within the solenoid,

${\displaystyle \displaystyle \Phi ={\frac {\mu _{0}\mu _{r}NiA}{l}},}$

from which it follows that the inductance of a solenoid is given by:

${\displaystyle \displaystyle L={\frac {\mu _{0}\mu _{r}N^{2}A}{l}}.}$

where N is squared because of the definition of inductance.

Since the permeability of ferromagnetic materials changes with applied magnetic flux, the inductance of a coil with a ferromagnetic core will generally vary with current.

### Inductance of a coaxial line

Let the inner conductor have radius ${\displaystyle r_{i}}$ and permeability ${\displaystyle \mu _{i}}$, let the dielectric between the inner and outer conductor have permeability ${\displaystyle \mu _{d}}$, and let the outer conductor have inner radius ${\displaystyle r_{o1}}$, outer radius ${\displaystyle r_{o2}}$, and permeability ${\displaystyle \mu _{o}}$. Assume that a DC current ${\displaystyle I}$ flows in opposite directions in the two conductors, with uniform current density. The magnetic field generated by these currents points in the azimuthal direction and is a function of radius ${\displaystyle r}$; it can be computed using Ampère's law:

{\displaystyle {\begin{aligned}0\leq r\leq r_{i}:B(r)&={\frac {\mu _{i}Ir}{2\pi r_{i}^{2}}}\\r_{i}\leq r\leq r_{o1}:B(r)&={\frac {\mu _{d}I}{2\pi r}}\\r_{o1}\leq r\leq r_{o2}:B(r)&={\frac {\mu _{o}I}{2\pi r}}\left({\frac {r_{o2}^{2}-r^{2}}{r_{o2}^{2}-r_{o1}^{2}}}\right)\end{aligned}}}

The flux per length ${\displaystyle l}$ in the region between the conductors can be computed by drawing a surface containing the axis:

${\displaystyle {\frac {d\phi _{d}}{dl}}=\int _{r_{i}}^{r_{o1}}B(r)dr={\frac {\mu _{d}I}{2\pi }}\ln {\frac {r_{o1}}{r_{i}}}}$

Inside the conductors, L can be computed by equating the energy stored in an inductor, ${\displaystyle {\frac {1}{2}}LI^{2}}$, with the energy stored in the magnetic field:

${\displaystyle {\frac {1}{2}}LI^{2}=\int _{V}{\frac {B^{2}}{2\mu }}dV}$

For a cylindrical geometry with no ${\displaystyle l}$ dependence, the energy per unit length is

${\displaystyle {\frac {1}{2}}L'I^{2}=\int _{r_{1}}^{r_{2}}{\frac {B^{2}}{2\mu }}2\pi r~dr}$

where ${\displaystyle L'}$ is the inductance per unit length. For the inner conductor, the integral on the right-hand-side is ${\displaystyle {\frac {\mu _{i}I^{2}}{16\pi }}}$; for the outer conductor it is

${\displaystyle {\frac {\mu _{o}I^{2}}{4\pi }}\left({\frac {r_{o2}^{2}}{r_{o2}^{2}-r_{o1}^{2}}}\right)^{2}\ln {\frac {r_{o2}}{r_{o1}}}-{\frac {\mu _{o}I^{2}}{8\pi }}\left({\frac {r_{o2}^{2}}{r_{o2}^{2}-r_{o1}^{2}}}\right)-{\frac {\mu _{o}I^{2}}{16\pi }}}$

Solving for ${\displaystyle L'}$ and summing the terms for each region together gives a total inductance per unit length of:

${\displaystyle L'={\frac {\mu _{i}}{8\pi }}+{\frac {\mu _{d}}{2\pi }}\ln {\frac {r_{o1}}{r_{i}}}+{\frac {\mu _{o}}{2\pi }}\left({\frac {r_{o2}^{2}}{r_{o2}^{2}-r_{o1}^{2}}}\right)^{2}\ln {\frac {r_{o2}}{r_{o1}}}-{\frac {\mu _{o}}{4\pi }}\left({\frac {r_{o2}^{2}}{r_{o2}^{2}-r_{o1}^{2}}}\right)-{\frac {\mu _{o}}{8\pi }}}$

However, for a typical coaxial line application, we are interested in passing (non-DC) signals at frequencies for which the resistive skin effect cannot be neglected. In most cases, the inner and outer conductor terms are negligible, in which case one may approximate

${\displaystyle L'={\frac {dL}{dl}}\approx {\frac {\mu _{d}}{2\pi }}\ln {\frac {r_{o1}}{r_{i}}}}$

## Phasor circuit analysis and impedance

If signals of current and voltage are sine, using phasors, the equivalent impedance of an inductance is given by:

${\displaystyle Z_{L}={\frac {V}{I}}=j\omega L\,}$

where

j is the imaginary unit,
L is the inductance,
ω = 2πf is the angular frequency,
f is the frequency and
ωL = XL is the inductive reactance.

## Nonlinear inductance

Many inductors make use of magnetic materials. These materials over a large enough range exhibit a nonlinear permeability with such effects as saturation. In turn, the saturation makes the resulting inductance a function of the applied current. Faraday's Law still holds but inductance is ambiguous and is different whether you are calculating circuit parameters or magnetic fluxes.

The secant or large-signal inductance is used in flux calculations. It is defined as:

${\displaystyle L_{s}(i)\ {\overset {\underset {\mathrm {def} }{}}{=}}\ {\frac {N\Phi }{i}}={\frac {\Lambda }{i}}}$

The differential or small-signal inductance, on the other hand, is used in calculating voltage. It is defined as:

${\displaystyle L_{d}(i)\ {\overset {\underset {\mathrm {def} }{}}{=}}\ {\frac {d(N\Phi )}{di}}={\frac {d\Lambda }{di}}}$

The circuit voltage for a nonlinear inductor is obtained via the differential inductance as shown by Faraday's Law and the chain rule of calculus.

${\displaystyle v(t)={\frac {d\Lambda }{dt}}={\frac {d\Lambda }{di}}{\frac {di}{dt}}=L_{d}(i){\frac {di}{dt}}}$

There are similar definitions for nonlinear mutual inductances.

## References

1. ^ a b Sears and Zemansky 1964:743
2. ^ Sears and Zemansksy 1964:671
3. ^ Sears and Zemansky 1964:671 – "The work of Oersted thus demonstrated that magnetic effects could be produced by moving electric charges, and that of Faraday and Henry that currents could be produced by moving magnets."
4. ^ Sears and Zemansky 1964:731 – "The direction of an induced current is such as to oppose the cause producing it".
5. ^ Heaviside, Oliver (1894). Electrical Papers. Macmillan and Company. p. 271.
6. ^ Glenn Elert. "The Physics Hypertextbook: Inductance". Retrieved 2016-07-30.
7. ^ Michael W. Davidson (1995–2008). "Molecular Expressions: Electricity and Magnetism Introduction: Inductance".
8. ^
9. ^ The kinetic energy of the drifting electrons is many orders of magnitude smaller than W, except for nanowires.
10. ^ Mahmood Nahvi; Joseph Edminister (2002). Schaum's outline of theory and problems of electric circuits. McGraw-Hill Professional. p. 338. ISBN 0-07-139307-2.
11. ^ e.g. Stephen C. Thierauf, High-speed Circuit Board Signal Integrity, p. 56, Artech House, 2004 ISBN 1580538460.
12. ^ Kim, Seok; Kim, Shin-Ae; Jung, Goeun; Kwon, Kee-Won; Chun, Jung-Hoon, "Design of a reliable broadband I/O employing T-coil", Journal of Semiconductor Technology and Science, vol. 9, iss. 4, pp. 198-204
13. ^ Radecki, A. et al., "Simultaneous 6-Gb/s Data and 10-mW Power Transmission Using Nested Clover Coils for Noncontact Memory Card," in IEEE Journal of Solid-State Circuits, vol. 47, no. 10, pp. 2484-2495, Oct. 2012. [1]
14. ^ Neumann, F. E. (1847). "Allgemeine Gesetze der inducirten elektrischen Ströme". Abhandlungen der Königlichen Akademie der Wissenschaften zu Berlin, aus dem Jahre 1845: 1–87.
15. ^ Dengler, R. (2016). "Self inductance of a wire loop as a curve integral". Advanced Electromagnetics. 5 (1): 1–8. Bibcode:2016AdEl....5....1D. doi:10.7716/aem.v5i1.331.
16. ^ Jackson, J. D. (1975). Classical Electrodynamics. Wiley. p. 262.
17. ^ Lorenz, L. (1879). "Über die Fortpflanzung der Elektrizität". Annalen der Physik. VII: 161–193. (The expression given is the inductance of a cylinder with a current around its surface). Bibcode:1879AnP...243..161L. doi:10.1002/andp.18792430602.
18. ^ Elliott, R. S. (1993). Electromagnetics. New York: IEEE Press. Note: The constant −3/2 in the result for a uniform current distribution is wrong.
19. ^ Rosa, E.B. (1908). "The Self and Mutual Inductances of Linear Conductors". Bulletin of the Bureau of Standards. 4 (2): 301–344. doi:10.6028/bulletin.088.
20. ^ D. Howard Dellinger; L. E. Whittmore; R. S. Ould (1924). "Radio Instruments and Measurements". NBS Circular. National Bureau of Standards. C74. Retrieved 2009-09-07.

## General references

• Frederick W. Grover (1952). Inductance Calculations. Dover Publications, New York.
• Griffiths, David J. (1998). Introduction to Electrodynamics (3rd ed.). Prentice Hall. ISBN 0-13-805326-X.
• Wangsness, Roald K. (1986). Electromagnetic Fields (2nd ed.). Wiley. ISBN 0-471-81186-6.
• Hughes, Edward. (2002). Electrical & Electronic Technology (8th ed.). Prentice Hall. ISBN 0-582-40519-X.
• Küpfmüller K., Einführung in die theoretische Elektrotechnik, Springer-Verlag, 1959.
• Heaviside O., Electrical Papers. Vol.1. – L.; N.Y.: Macmillan, 1892, p. 429-560.
• Fritz Langford-Smith, editor (1953). Radiotron Designer's Handbook, 4th Edition, Amalgamated Wireless Valve Company Pty., Ltd. Chapter 10, "Calculation of Inductance" (pp. 429–448), includes a wealth of formulas and nomographs for coils, solenoids, and mutual inductance.
• F. W. Sears and M. W. Zemansky 1964 University Physics: Third Edition (Complete Volume), Addison-Wesley Publishing Company, Inc. Reading MA, LCCC 63-15265 (no ISBN).